The explosive growth in mobile electronic devices such as smartphones and tablets creates an increasing need in the art for compact and efficient switching power converters so that users may recharge these devices. A flyback switching power converter is typically provided with a mobile device as its transformer provides safe isolation from AC household current. This isolation introduces a problem in that the power switching occurs at the primary side of the transformer but the load is on the secondary side. The power switching modulation for a flyback converter requires knowledge of the output voltage on the secondary side of the transformer. Such feedback can be obtained through opto-isolators bridging from the secondary side to the primary side but this adds to cost and control complexity. Thus, primary-only feedback techniques have been developed that use a feedback voltage at an auxiliary winding on the primary side, which reflects the output voltage on the secondary side in each switching cycle.
In an example flyback converter 100 that utilizes primary-only feedback, as shown in FIG. 1, a feedback voltage V_FB signal is provided by sensing a voltage at a primary auxiliary winding of a transformer T1. FIG. 2A-C shows waveforms associated with flyback converter 100 of FIG. 1. FIG. 2A shows a waveform for an ON and OFF state of a power switch S1, FIG. 2B shows a waveform for the feedback voltage at the primary auxiliary winding, and FIG. 2C shows a waveform for secondary current (current in the secondary winding of the transformer).
When power switch S1 is placed in the ON state, primary current starts to flow through the primary winding of transformer T1. Since a diode D1 coupled to the secondary winding of transformer T1 is reverse biased, the secondary current is zero, causing energy to be stored in the core of transformer T1.
When power switch S1 is switched to the OFF state, diode D1 becomes forward biased, the secondary current pulses high, and the energy stored in transformer T1 starts to be delivered to the secondary. This can be observed by the forward current through diode D1. As energy is delivered to the secondary, current through diode D1 decreases linearly. Once all of the energy stored in transformer T1 has been expended, the secondary current through diode D1 ramps down and reaches 0 A (amperes) as power is delivered to the load. This point is commonly referred to as the transformer reset point. Further, the delay between the power switch off time and the secondary current ramping to zero is denoted as the transformer reset time T_RST.
During the period between the turn OFF of power switch S1 and the transformer reset point, the relationship between the feedback voltage across the auxiliary winding and the output voltage can be expressed by “V_FB=((V_OUT)−(V_D1+V_IR))/N,” where V_FB=feedback voltage across the auxiliary winding, V_OUT=output voltage, N=auxiliary winding to secondary winding turns ratio, V_D1=forward voltage drop across diode D1, and V_IR=voltage drop due to secondary I×R losses. Therefore, to achieve an accurate representation of output voltage V_OUT using feedback voltage V_FB, it is desirable to obtain feedback voltage V_FB by sampling the auxiliary winding voltage at the transformer reset point. At this point, V_IR losses equal 0 V (Volts).
Although the ideal sampling point for feedback voltage V_FB is the transformer reset point, special care should be taken to insure the feedback voltage V_FB sampling point does not occur after the transformer reset point, as the voltage across the auxiliary winding then does not represent output voltage V_OUT. Therefore, the sampling point is programmed to occur when secondary current is at a very low point, such as just prior to the transformer reset point where V_IR =˜0 V. For example, if the sampling point is at a fixed time set-back from the transformer reset point on a cycle-by-cycle basis, V_IR will have a constant value.
Another reason to insure the feedback voltage V_FB sampling point occurs when the secondary current is at a low and fixed value is to insure forward voltage drop V_D1 across diode D1 is a constant value on a cycle-by-cycle basis. The forward voltage drop of commonly used output rectifiers such as diode D1 are typically in the range of 0.2 V at the current levels where the feedback voltage V_FB signal is obtained. The feedback voltage V_FB signal at the transformer reset time T_RST is proportional to output voltage V_OUT (based upon the turn ratio in the transformer and other factors) offset by forward voltage drop V_D1. Primary-only feedback techniques use this feedback voltage V_FB signal to efficiently modulate the power switching and thus modulate the output voltage V_OUT, such as to 5 V.
However, having a fixed output voltage such as 5 V is problematic for fast charging of modern devices. In particular, it is conventional for a switching power converter to couple to the device being charged through a standard interface such as a Universal Serial Bus (USB) interface. The USB interface includes a differential pair of signals (D+ and D−) for signaling and also provides power and ground. With regard to the delivery of power, a USB cable can only support a certain amount of current. For example, the USB 2.0 standard allows for a maximum output current of 500 mA, whereas the USB 3.0 standard can support a maximum output current of 900 mA. Traditionally, the delivery of power through a USB cable used a voltage of 5 V. But modern mobile device batteries have relatively large storage capacities such as several thousand mA (milliamps). The charging of such batteries, even at the increased output currents allowed in the USB 3.0 standard, will thus be delayed if the power is delivered using a 5 V power supply voltage. This is particularly true in that the switching power converter, the cable, and the receiving device all present a resistance to the output current.
To enable a rapid charge mode in light of the output current limitations and associated losses from device resistances, it is now becoming conventional to use markedly higher output voltages over the USB cable. For example, rather than use the conventional USB output voltage of 5 V, multi-level power converters configured to provide a regulated output voltage at multiple regulation levels (e.g., multiple settings) have been developed, such as the “quick charge” power converters that are used for USB-based portable equipment. A multi-level power converter may support rapid charge modes using output voltages of 9 V, 12 V, or even 19 V. In addition to the standard 5 V setting, the multi-level power converter may thus provide a higher regulated output voltage (e.g. 9 V, 12 V, or 19 V) depending on the portable device that is connected to the multi-level power converter. The increased output voltages allow the multi-level power converter to deliver more power over the USB cable without exceeding the maximum output current limitations. As many legacy devices can only accept the standard 5V from a USB cable, the multi-level power converter will engage in an enumeration process with the device being charged to determine if the higher output voltages are supported. This enumeration may occur over the differential D+ and D− pins. Through the enumeration, the multi-level power converter and the enumerated device may change the USB output voltage to an increased level that is supported by the enumerated device. The result is considerably reduced charging time, which leads to greater user satisfaction.
Although primary-only feedback techniques advantageously eliminate the need for secondary-side regulators and opto-isolators, problems have arisen in their implementation. For example, there is a voltage offset that must be compensated due to forward voltage drop V_D1 of diode D1. An example single-voltage output power converter 300 is shown in FIG. 3 in which the reference voltage is offset to compensate for the forward voltage drop V_D1. For example, if the desired output voltage is 5 V, and a reference voltage of 5 V provides an output voltage of 4.8V due to the forward voltage drop of diode D1, the reference voltage V_REF is offset to 5.2 V to obtain the desired output voltage of 5 V (assuming a turns ratio N of 1). However, this creates a problem for multi-level power converters as scaling the reference voltage V_REF to obtain a higher desired output voltage results in an inaccurate output voltage.
For example, a multi-level power converter may have three output voltage settings, 5 V, 12 V, and 19 V. The multi-level power converter uses an initial reference voltage V_REF0 (e.g., 5.2 V) that is offset from the initial output voltage V_OUT0 of 5 V as in FIG. 3. Then, for the other desired output voltages V_OUT1, 12 V and 19 V, the initial reference voltage V_REF0 is scaled by a ratio of the desired output voltage V_OUT1 and the initial output voltage V_OUT0. This is represented by the formula:
      V    REFn    =            V              REF        ⁢                                  ⁢        0              ×                  V                  OUT          ⁢                                          ⁢          1                            V                  OUT          ⁢                                          ⁢          0                    The initial reference voltage V_REF0 is scaled by 12/5 to obtain the output voltage of 12 V and by 19/5 to obtain the output voltage of 19 V. However, as mentioned above, since there is a voltage offset between the initial reference voltage V_REF0 and the initial output voltage V_OUT0 mainly due to the forward voltage drop V_D1 across diode D1, scaling the initial reference voltage V_REF0 results in the inaccurate setting of the 12 V and 19 V levels. This is because the offset does not scale with the output voltage setting since the forward voltage drop V_D1 across the output diode D1 remains constant regardless of the output voltage. This is shown by Table 1.
TABLE 1DesiredV_OUTScaling RatioV_REFV_OUT+5.0V1+5.2 V (Set at final test)+5.0V+12.0V2.4 (12/5)+12.48 V+12.28V+19.0V3.8 (19/5)+19.76 V+19.56V
To adjust an inaccurate output voltage, conventional multi-level power converters trim the voltage at the auxiliary winding using feedback voltage dividers. As shown in FIG. 4, an example multi-level power converter 400 includes a feedback voltage divider formed by resistors R1 and R2 to provide a trimmed feedback voltage V_FB. Multi-level power converter 400 also includes a resistor R3 and a rapid-charge mode switch S2 coupled between a feedback voltage input of a comparator U2 and ground. If rapid-mode switch S1 is switched on, resistors R3 and R2 are coupled in parallel such that the resistance between the feedback voltage input and ground drops. Feedback voltage V_FB will thus drop when switch S1 is switched on, which causes an error signal from comparator U2 to increase and, in response, output voltage V_OUT will increase. Conversely, if the rapid-mode switch S2 is switched off, output voltage V_OUT will drop in response to the increase in feedback voltage V_FB. A mode control circuit U3 controls the rapid-charge mode switch S2 to select between the output voltage settings. The resistor values of resistors R1, R2, and R3 that results in the desired output voltage may be determined and set through trial and error testing. Depending on the number of output voltage settings, conventional multi-level power converters may include multiple resistors and switches so that the auxiliary winding voltage is trimmed by a different voltage divider or combination of resistors for each voltage setting. The more settings a multi-level power converter has, the more resistors and switches are required, as a separate voltage divider or combination of resistors is required for each voltage setting. When a desired output voltage is determined, the switches are used to couple the auxiliary winding to a voltage divider corresponding to that output voltage.
However, trimming the auxiliary winding voltage and/or the reference voltage using multiple resistors and switches increases the number of components required for power converters, which increases the cost of such power converters and also increases the chances of component failure. Also, trimming the auxiliary winding voltage and/or the reference voltage requires costly post-production tests. Moreover, the number of output voltage settings that may be provided by trimming of feedback voltage V_FB is limited since a different combination of resistors or voltage dividers is required for each output voltage setting. Indeed, conventional multi-level power converters cannot vary an output voltage on a continuum, but typically only can support three to five pre-set voltages.
Accordingly, there is a need in the art for improved flyback converters and flyback control techniques to provide an output voltage at various regulation levels.